In order to convey high frequency signals, a so-called micro strip technology implemented on printed circuit is, for example, used. Printed conductive tracts are produced on faces of the printed circuit. These faces may be external or internal and may be separated by one or more conductive planes or ground planes. The micro strip lines have particular dimensions such that, once associated with the conductive planes, they form impedance-matched lines. This matching makes it possible to provide a certain transparency of the line with regard to the signal conveyed. In other words, the aim is to minimize the electrical power losses of the signal along the line.
To convey the high frequency signals outside the printed circuit, coaxial connectors for connecting the micro strip lines to coaxial cables, whose impedances are also matched, are, for example, used. There are connectors intended for surface-mounting. This type of mounting does not require any metallized hole for a contact of the connector to be inserted into. More specifically, these connectors are connected to the printed circuit by bonding flat areas of the connector to external lands of the printed circuit. The connection may be made by means of a conductive glue or a solder paste interposed between the flat areas and the corresponding lands. The printed circuit on which the components are placed, such as the connectors, is immersed in a tank in which a hot fluid, generally in vapour phase, is used to solder the components.
FIG. 1 illustrates one embodiment of a printed circuit 10 intended to receive a connector 11 on one of its external faces 12. This embodiment was implemented by the applicant without being disclosed. On this face, a transmission line 13, produced in the form of a micro strip line, comprises a rectilinear part 14 terminated by a circular bump contact 15, the diameter of which is greater than the width of the rectilinear part 14. The width of the rectilinear part 14 is defined so that the transmission line has a determined impedance, for example 50 ohms. This impedance depends on the composition and the dimensions of the printed circuit 10. For example, the printed circuit comprises an insulating substrate of relative permittivity εr equal to 3.38, the substrate separating the external face 12 from an internal ground plane situated facing the transmission line 13. The internal ground plane is not visible in FIG. 1. To obtain an impedance of 50 ohms, a width for the rectilinear part 14 of 450 μm and a thickness of the substrate of 203 μm are, for example, selected. As for the diameter of the bump contact 15, this is imposed by the diameter of the flat area of the core of the connector 11. In FIG. 1, the core cannot be seen. It is concealed by the body 16 of the connector 11. The body 16 is connected by flat areas 17 to a ground plane 18 produced on the external face 12 and surrounding the transmission line 13. In the example shown, the diameter of the bump contact 15 is 1.7 mm. Metallized holes or vias 19 are used to connect the ground plane 18 to the internal ground plane. The ground plane 18 is positioned at a distance from the transmission line 13 so that its interaction with the transmission line 13 does not modify, or modifies only negligibly, the impedance of the transmission line 13 while providing shielding for the latter. The edges of the ground plane 18 are, for example, situated at a distance d of 500 μm from the transmission line 13. The ground plane 18 notably allows for easier grounding for the surface-mounted components, in particular for the connector 11. The ground plane 18 also provides strong electrical insulation between the various patterns printed on the external face 12.
FIGS. 1a and 1b represent enlarged views of the transmission line 13 in its plane. FIG. 1a represents a portion of the rectilinear part 14. All along this part, the distance d is constant. FIG. 1b represents the transmission line 13 at its bump contact 15. The distance d, defined for the rectilinear part 14, is maintained around the bump contact 15.
Connecting the connector 11 with a multilayer structure, as illustrated in FIG. 1, produces spurious electrical effects, which in particular break the transparency of the connector with regard to the signal conveyed. This transparency is above all degraded at high frequency. These effects, notably linked to the intrinsic capacitive nature of the bump contact 15, then result in mismatching and greater or lesser insertion losses on the signal.
FIG. 2 represents the matching level S11 of the example represented in FIG. 1 expressed in dB according to the frequency of the signal conveyed by the transmission line 13 and the connector 11. The matching level illustrates the reflected electrical power. Still for this example, FIG. 3 represents the insertion level S21 or insertion losses expressed in dB according to the frequency of the signal. The insertion level illustrates the electrical power losses in transmission at the connection between the transmission line 13 and the connector 11.
It is found, in this case, that this connection exhibits a matching level of −6 dB for a useful working frequency of 9.3 GHz, and insertion losses of 1.5 dB for this same frequency. Given these results, the connection is far from optimal.
To improve the transparency of this connection, the applicant has proposed correcting this mismatching. An exemplary embodiment of this correction is represented in FIG. 4. The transmission line 13 comprises, in the vicinity of the bump contact 15, an inductive line section 20 and a capacitive line section 21 that make it possible to produce a low-pass type filtering element cooperating with the impedance added by the transition between the connector 11 and the bump contact 15. The inductive line section 20 is formed by a micro strip line, the width of which is less than that of the rectilinear part 14. The capacitive line section 21 is formed by a square-shaped bump contact, the side dimension of which is greater than the width of the rectilinear part 14. The inductive line section 20 is positioned between the bump contact 15 and the capacitive line section 21. In a first approach, the various electrical elements of the transition, associated with impedance values at a given frequency, form a π-form low-pass filter cell comprising an inductance, the inductive line section 20, positioned between two capacitances, the capacitive line section 21 and the bump contact 15. The dimension of the bump contact 15 is imposed by the connector 11 and the dimensions of the impedances 20 and 21 are matched to obtain the best transparency for the transition at the frequency concerned according to the characteristics of the printed circuit 10, thickness and permittivity of the substrate.
FIG. 5 represents the matching level S11 of the example represented in FIG. 4 expressed in dB according to the frequency of the signal conveyed by the transmission line 13 and the connector 11. Still for this example, FIG. 6 represents the insertion losses expressed in dB according to the frequency of the signal. It is found, in this case, that this transition exhibits a matching level of approximately −18 dB for a useful working frequency of 9.3 GHz, and insertion losses of 0.27 dB for this same frequency. It is found that the correction made significantly improves the transparency of the transition. However, the results obtained exhibit a relatively narrow-band nature around the working frequency. It is therefore necessary to modify the design of the printed circuit for any change of working frequency, however small. Furthermore, the quality of the correction obtained in this way is highly sensitive to the production tolerances of the printed circuit and the positioning tolerances of the connector on the printed circuit. In other words, there is a risk of obtaining significant random effects on the quality of the correction between two printed circuits produced according to the same design.